Wireless repeater with fir based channel equalizer

ABSTRACT

This invention presents a repeater enhanced MU-MIMO wireless communication system comprising a BS, a plural of repeaters, and a plural of UEs, where a repeater estimates the channel between itself and its upper communication node in the system, a repeater computes equalization coefficients based on the estimation of the channel coefficients, and a repeater applies the equalization coefficients to reduce the channel delay spread or increase the coherence bandwidth of the channel between communication nodes containing the BS, the UEs, or the repeaters.

This application claims the benefit of U.S. Provisional Application No.62/157,471, filed on May 6, 2015.

FIELD OF INVENTION

This invention relates generally to novel relay designs to increasecoherence bandwidth of wireless channels with Finite Impulse Response(FIR) filters in wireless systems.

BACKGROUND

With the proliferation of mobile applications, there is an increasingdemand for higher throughput of wireless systems at a staggering pace.Given the fact that the limited spectrum under 6-GHz is already crowded,millimeter Wave (mmWave) has emerged as a promising technology of futureFifth-Generation (5G) wireless systems [1].

Properly designed repeaters can play an important role in wirelesssystems. In Wireless Fidelity (WiFi) or Long Term Evolution (LTE)systems, repeaters are used to extend the coverage range. For mmWave,the role of repeaters is fundamentally different. Given the strong radiopropagation directivity and large reflection loss of mmWave signals,repeaters are essential for seamless coverage [2]. Note that repeaterscan be divided into two categories: Amplify-and-Forward Repeater (AFR)and Decode-and-Forward Repeater (DFR). Since the DFR introducesconsiderable propagation delay especially for multi-hop repeaterscenario, AFR enhanced wireless systems offer advantages over DFRs. Theenergy efficiency of repeaters was studied in [3]. The problem ofminimizing the number of repeaters and maximizing network utilities wasstudied in [4]. In [5], an iterative algorithm is developed for jointlydesigning the Receive/Transmit (Rx/Tx) Radio Frequency (RF)/basebandprocessors. It was demonstrated that multi-hop repeaters can greatlyimprove the connectivity versus single hop mmWave transmission in [6].

There have been previous inventions on utilizing repeaters in wirelesssystems. However, little attention has been paid to the impact ofrepeaters on wireless channel coherence bandwidth. Coherence bandwidthmeans all the sub-carriers within it share the similar channelcharacteristics, so that channel estimation only needs to be performedonce for all the sub-carriers. In [7], it is demonstrated that multi-hoprepeaters will make the channel coherence bandwidth narrower, but nomethods were proposed to combat this issue of narrowed coherencebandwidth. Narrower coherence bandwidth means more resources (e.g.,pilot spectrum and computation) have to be spent on channel estimation.

One embodiment of this invention is an innovative repeater design withan FIR-based channel equalizer to increase the coherence bandwidth ofwireless systems. Based on the fact that the channels between repeatersare slow varying due to the repeaters being static (i.e., not moving),and thus have long coherence time, each repeater can adaptively equalizethe channel using recently obtained channel estimates. With theequalized channels by this novel design, the energy sensitive UserEquipments (UEs) can spend much less resources on channel estimation.For example, in an mmWave system with 4 repeaters and 100 UEs, all UEscan save half the resources in channel estimation, if the 4 repeatersare equipped with the FIR filters proposed in this invention.

Another embodiment of this invention is to equalize channels withbeamforming with multiple antennas at transmitter or receiver. If thenumber of transmitter antennas is N_(t) and the number of receiverantennas is N_(r), the repeater would need N_(t) N_(r) FIR filters. Toreduce the complexity especially for systems with large numbers ofantennas, this invention describes a method to first perform transmitteror receiver beamforming and then equalize the channels at each receiverantennas. Then, only N_(r) FIR filters are needed for the repeater withN_(r) antennas.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 shows the system model of repeaters.

FIG. 2 shows the system block of a single-antenna repeater.

FIG. 3 illustrates the structure of one equalization FIR filter.

FIG. 4 shows a time-domain example of equalizer.

FIG. 5 shows the simulation results of coherence bandwidth withdifferent layers of repeaters.

FIG. 6 shows the simulation results of Bit Error Rate (BER) withdifferent layers of repeaters.

FIG. 7 shows the simulation results for different subcarrier-groupingstrategies.

FIG. 8 shows the diagram of Multiple-Input-Multiple-Output (MIMO)repeaters.

FIG. 9 shows the functional block diagram of an MIMO repeater withtransmitter side beamforming algorithm.

FIG. 10 shows the functional block diagram of an MIMO repeater withreceiver side beamforming algorithm.

FIG. 11 shows the block chart of receiver side Zero-Forcing (ZF)algorithm as an example.

FIG. 12 shows the simulation results of coherence bandwidth for MIMOrepeaters.

FIG. 13 shows the flow chart of FIR parameter estimation without channelfeedback.

FIG. 14 shows the flow chart of FIR parameter estimation with channelfeedback.

DETAILED DESCRIPTION

Reference may now be made to the drawings wherein like numerals refer tolike parts throughout. Exemplary embodiments of the invention may now bedescribed. The exemplary embodiments are provided to illustrate aspectsof the invention and should not be construed as limiting the scope ofthe invention. When the exemplary embodiments are described withreference to block diagrams or flowcharts, each block may represent amethod step or an apparatus element for performing the method step.Depending upon the implementation, the corresponding apparatus elementmay be configured in hardware, software, firmware or combinationsthereof.

Equalization Based on FIR Filter for One Pair of Transmitter-Receiver

The system model of the repeaters is shown in FIG. 1, where the directpath between the Base Station (BS) 1 and the UE 2 is blocked byobstacles 3, thus, the wireless signal propagates through the repeaters4 to the UE.

Motivation of Equalization to Increasing Coherence Bandwidth: Channelfrequency selectivity can be characterized by coherence bandwidth. Sincesub-carriers within the coherence bandwidth have the similar channel,the UE or the BS only needs to estimate the channel once for all thesub-carriers within the coherence bandwidth. If the coherence bandwidthis wider, the system could spend less resource (e.g., pilot andcomputation) in channel estimation. As an example of LTE systems, thereare totally 1200 sub-carriers. If the coherence bandwidth is 48sub-carriers, the 1200 sub-carriers are divided into 25 groups, and eachgroup only needs to conduct channel estimation once.

Assume that the transmitted signal is x(t), and the received signal y(t)through the wireless channel denoted by h(t) is then denoted as

y(t)=x(t)*h(t)=∫_(τ) ^(∞) x(t−τ)h(τ)dτ,   (1)

where * denotes convolution.

Let h_(k)(t) denote the impulse response of the kth, k=1, 2, . . . K,hop, where the first hop begins from the BS, and the last hop ends atthe UE. If there is no repeater during signal propagation, then K=1.Hence, the received signal through K hop of repeaters is given as

y(t)=x(t)*h ₁(t)*h ₂(t)* . . . *h _(K)(t)=x(t)*h ^(r)(t),   (2)

where h^(r)(t)=Π_(k=1) ^(K)h_(k)(t) denotes the overall channel impulseresponse through K−1 repeaters.

One embodiment of this invention is the repeater shown in FIG. 2, whichconsists of two antennas 5, two bandpass filters 6, an FIR filter 7, andan amplifier 8 for each direction, i.e., the downlink direction and theuplink direction. Note that the FIR filter may be placed after theamplifier. The FIR filter is designed to equalize the wireless channels.The system block of the FIR filter is illustrated in FIG. 3, whichconsists of L−1 delayers 9, L multipliers 10, and one adder 11, wherethere are L taps. The received signal y(m) 12 is first passed to thedelayers. Then, the filter coefficients w(0), . . . , w(L−1) 13 areapplied to the delayed signals by the multipliers. Finally, the filteredsignals are summed by the adder to generate the output signal{circumflex over (x)}(m) 14. Note that the repeater might have othercomponents to have other functions. For example, an attenuator and aphase shifter can be added into the repeater to create reciprocal uplinkand downlink inside paths, as invented in our patent applicationPCT/US16/13744.

Assume that the impulse response of FIR filter on the ith repeater isw_(i)(t), i=1, . . . , K−1, then the final received signal can bewritten as a discrete time form

y(m)=x(m)*Π_(i=1) ^(K−1)[h _(i)(m)*w _(i)(m)]*h _(K)(m),   (3)

Where y(m)=y(mT_(s)) with T_(s) being the sampling rate. Note that thedestination of the last hop h_(K)(m) is the receiver, so there is nocorresponding equalizer.

Another embodiment of this invention is the method to calculate thevalues of w_(i) (m) shown as follows. At the ith receiver, it estimatesthe channel h_(i), then calculates w_(i) to equalize it. Let x_(i)^(P)(m) denotes the training pilot used to estimate h_(i) The reasonthat the channel h_(i) is frequency selective is that the receivedsignal at the ith repeater y′_(i)(m)=x_(i) ^(p)(m)*h_(i)(m) consistssome delayed replica of previous data x_(i) ^(p)(m−1), x_(i) ^(p)(m−2),. . . . If y(m) is not corrupted by previous data, then y′_(i)(m)=x_(i)^(p)(m)h_(i)(m) and the channel is flat. Therefore, the problem isessentially designing a filter w_(i) so that the output {circumflex over(x)}_(i) ^(p)(m)=y_(i)′(m)*w_(i)(m) is close to x_(i) ^(p)(m). Withoutloss of generality, assume that the pilot signal x_(i) ^(p)(m) is thesame for all the repeaters. For simplicity, let {circumflex over (x)}(m)and x(m) denote {circumflex over (x)}_(i) ^(p)(m) and x_(i) ^(p)(m),respectively. Then, the goal of equalization is to choose w_(i) tominimize E(|{circumflex over (x)}(m)×(m)|²)

Assume that w_(i) has L taps, then,

{circumflex over (x)}(m)= w _(i)(m)^(H) y′ _(i)(m),   (4)

where

y′ _(i)(m)=[y′ _(i)(m−L+1),y′ _(i)(m−L+2), . . . , y′ _(i)(m)]^(T)   (5)

and

w _(i)(m)=[w* _(i)(L−1),w* _(i)(L−2), . . . , w* _(L)(0)]^(T).   (6)

To estimate the channel, the repeater does not need to decode thesignals, but need to do analog-to-digital sampling. Like other RFcomponents on the repeater (such as bandpass filter and amplifier), theFIR filter will introduce additional delay, but it is fixed and themaximum delay is the length of taps and can be designed to stay withinthe delay tolerance of the total channel, e.g., keeping the cyclicprefix under a maximum value. Therefore, the problem can be defined asthe following:

minE(|{circumflex over (x)}(m)−x(m)|²)=min _(w) _(i) _((m)) E[|w_(i)(m)^(H) y′ _(i)(m)−x(m)|²].   (7)

Noticing the above is essentially a Minimum Mean Square Estimation(MMSE) problem, and the optimum solution satisfies that the estimationerror w _(i)(m)^(H) y′_(i)(m)−x(m) is orthogonal to the observationsy′_(i)(m)

E{y′ _(i)(m−1)[ w _(i)(m)^(H) y′ _(i)(m)−x(m)]*}=0,   (8)

with l=0, L−1. Then, the optimum solution is

w _(i)(m)=Cov[ y′ _(i)(m), y′ _(i)(m)]Cov[ y′ _(i)(m),x(m)]  (9)

where Cov denotes covariance.

Optionally, to guarantee that the input and output (of the FIR filter)signals have the same power, w _(i)(m) can be normalized so that w_(i)(m)^(H) w _(i)(m)=1.

With the knowledge of x(m), the repeaters can compute the optimumw_(i)(m) based on the received signal y′_(i)(m). Since the channel ofeach hop h_(i)(m) is equalized, the overall channel of repeater hopsh₁(m)*h₂(m)* . . . *h_(K−1)(m) is equalized. Note that the repeaters canbe trained based on the existing downlink/uplink pilot signals. Sincethe repeaters are static, the channel h_(i) has a long coherence timeand this training can be done much less frequently than thedownlink/uplink channel estimation. In addition, the repeaters arealways less energy-sensitive than UEs. With this novel design, theenergy-sensitive UEs can spend much less resources for channelestimation.

Note that it is not required that every repeater has an FIR filter forequalization. For example, the system can use the kth repeater toequalize the channel from transmitter to it through the k−1 repeaters.

FIG. 4 shows one example of the FIR equalizer. The blue solid linedenotes the time domain h_(i) generated using channel model in LTEstandards [8]. The red dotted line is the channel after the FIRequalizer h_(i)*w_(i). As shown in the figure, the channel impulseresponse with the equalizer is close to zero for m≥3, so that thechannel is less disperse, leading to a wider coherence bandwidth.

The following simulation results show impact of the FIR equalizer oncoherence bandwidth and BER. In the simulation, the delay values of theN taps of channel responses are generated based on the Poissondistribution. As verified by the mmWave campaign that “The distributionof power among the path clusters is well modeled via a 3GPP model” [2],the power levels of the N taps of channel responses are generated basedon the exponential distribution [8]. Assume that the sampling rate atreceiver/repeaters is 3.072 GHz, and the channel bandwidth is 2 GHzdivided into 1200 subcarriers. Note that the values are chosen toachieve the same ratio of sampling rate over bandwidth as LTE [8]. Themaximum tap of an FIR filter is L=50, i.e., 16.3 ns.

Coherence Bandwidth: FIG. 5 shows the comparison between coherencebandwidth with and without equalizer for different numbers of hops. TheSignal-to-Noise (SNR) is set to be 30 dB. The coherence bandwidth is themaximum separation that the correlation is above a value η. For example,if η=0.8, the coherence bandwidth with “Repeater hop=0” is about 52subcarriers. The results show that the coherence bandwidth decreases asthe number of hops increases, and the proposed equalizer cansignificantly increase the coherence bandwidth, e.g., for “Repeaterhop=4”, the coherence bandwidth with equalizer is about 40 subcarriers,which is double that of about 20 subcarriers without equalizer.

BER: FIG. 6 shows the BER with different numbers of hops. Thefrequency-domain Binary Phase-Shift Keying (BPSK) OrthogonalFrequency-Division Multiplexing (OFDM) signals are first transformed tothe time domain, and then pass through the frequency selective channel.At the receiver, the channel of every 48 sub-carriers is estimated once,and the estimated channel is used to decode the BPSK signals of the 48sub-carriers. To show the effect of channel selectivity, the channelestimation is assumed to be perfect, therefore if the 48 sub-carriershave the same channel, the BER would be 0. For simplicity, no forwarderror correction is applied. Let G denote the number of sub-carriersthat use the same channel estimation for decoding. FIG. 7 shows the BERsfor different sub-carrier grouping methods with different values of G.The x-axis denotes the value of G and the y-axis is the BER. It showsthat more hops or a larger value of G leads to a higher BER. It alsoshows that the proposed equalizer is able to significant decrease BER,e.g., the BER with equalizer is about half that without equalizer for 4hops with G=48.

Equalization Based on FIR Filter for Multiple Transmitters-Receivers

If there are multiple transmitter or receiver antennas, each antenna ona repeater 4 will receive signals from multiple antennas on thetransmitter 15, which could be a BS, a UE, or anther repeater, as shownin FIG. 8. One embodiment of this invention is a method to firstcalculate the beamforming/precoding matrix at the transmitter, (if thenumber of transmitter antennas is larger than the number of receiverantennas) or to compute the beamforming/detection matrix at the receiver(if the number of transmitter antennas is equal or smaller than thenumber of receiver antennas), using methods such as ZF or MMSE, and theneach receiver equalizes the overall channels. FIG. 9 shows the systemlevel block diagram of transmitter side beamforming where thetransmitted symbol vector s 16 is firstly precoded by a beamformingmatrix 17 at the BS before being transmitted to the repeater, and FIG.10 shows the block diagram of receiver side beamforming where thereceived signal vector after the bandpass filters is multiplied by abeamforming matrix 18 at the repeater before being passed to the FIRfilters.

If the number of transmitter antennas is larger than the number ofreceiver antennas (repeaters in the first layer, i.e., the BS or the UE,has more antennas than repeaters), the transmitter side beamforming isneeded. If the number of receiver antennas is equal or larger than thenumber of transmitter antennas, the receiver side beamforming isrequired to separate the data streams. If the transmitter has moreantennas than the receiver, the transmitter needs to know the channelwhich can be obtained through uplink channel estimation (based onchannel reciprocity) or channel estimation feedback from receivers totransmitters. Otherwise, only the receivers need to know the channel toseparate data stream, and the channel can he estimated by downlink pilottransmission.

As shown in FIG. 8, there are N_(t) transmitter antennas, and N_(r)receiver antennas. Assume that the transmitter has more antennas thanthe receiver. Then, the transmitter sends N_(r) data s_(i), i=1, . . . ,N_(r), to the N_(r) receiver antennas simultaneously where s_(i) is thedesired signal for the ith antenna at the receiver, while others areinterferences.

The precoding matrix is defined as x=Ps, where x=[x₁, x₂, . . . , x_(N)_(t) ]^(T) is the data vector at the N_(t) transmitter antennas, s=[s₁,s₂, . . . , s_(N) _(r) ]^(T) is the data to be transmitted, and P is theprecoding matrix, with p_(i,j) being the coefficient of mapping the jthdata to the ith transmitter antenna. Let the jth column of P bep_(j)=[p_(1,j), p_(2,j), . . . , p_(N) _(t) _(,j)]^(T), then thereceived signal at the jth receiver antenna is y_(j)=p_(j)^(T)h_(j)s_(i) where h_(j)=[h_(1,j), h_(2,j), . . . , h_(N) _(t)_(,j)]^(T) with h_(i,j) being the channel from the ith transmitterantenna to the jth receiver antenna.

One procedure to compute the precoding matrix is described as follows.In the Time-Division Duplex (TDD) scenario, the N_(r) receiver antennassend pilot signals to the N_(t) transmitter antennas, then thetransmitter estimates the downlink channels based on channelreciprocity. In the Frequency-Division Duplex (FDD) version, the N_(t)transmitter antennas send pilot signals, and the N_(r) receiver antennasestimate the channels and feed hack the channel estimates to thetransmitter. Based on the channel estimation feedback, the optimumbeamforming matrix can be computed, e.g., using ZF, MMSE, or othermethods.

Then, the jth receiver antenna receives data through the equivalentchannel h _(j)=p_(j) ^(T)h_(j). The equalizer is then to equalize h _(j)based on the same method in the single pair of antennas scenariodescribed in the previous section.

One embodiment of this invention is the transmitter or receiverbeamforming algorithms to separate the data streams, so that theequalizer filter coefficients can be calculated for each data stream.One embodiment of this invention is that if the number of receiverantennas is equal or larger than the number of transmitter antennas, thereceiver can separate the data streams through data processing such asZF, MMSE, or other methods. In FIG. 11, the flowchart of receiver sideZF is shown as an example. After channel estimation, based on theestimated channel matrix, the ZF matrix P can be used to separate datastreams. In this embodiment, the transmitter does not need to know thechannel information, and the receiver antennas can estimate the channelbased on training pilots from the transmitter. Specifically, first, eachtransmitter antenna sends out pilot signals 19. Then, each receiverantenna estimates the channel between it and each transmitter antenna20. Next, based on the estimated channel matrix H, the ZF processing isy=PH where P=(H^(H)H)⁻¹H^(H) 21.

One difference from the single pair of antennas system is that the jthreceiver antenna might receive interference (transmitted data other thans_(i)). If the precoding matrix is not perfectly calculated, then, theestimation of h _(j) might not be accurate. In the following simulation,we assume imperfect beamforming with 10 dBSignal-to-Interference-plus-Noise Ratio (SINR) at the receiver.Simulation results in FIG. 12 show that the proposed algorithm alsoachieves good performance, i.e., equalization doubles the coherencebandwidth.

Procedure of Equalization in Wireless Systems

This section describes the procedure of the channel equalization withrepeaters in wireless systems, which includes the FIR parameterestimation, channel feedback, and UE channel estimation.

As described in previous sections, it is important to estimate thechannels between repeater antennas and transmitter antennas, so that theoptimum weighting of FIR filters can be calculated. As there might bemany repeaters on the same hop, we define the repeaters receivingsignals y_(i) (m) as the repeaters on the ith layer. Note that thechannel estimation can be obtained either by direct downlink channelestimation (the repeater on the ith layer equalizes channels between the(i−1)th layer and the ith layer) or through channel feedback (therepeater on the ith layer equalizes channels between the ith layer andthe (i+1)th layer). Note that the 0th layer is the BS for the downlinkand the UE for the uplink. FIG. 13 shows the process of direct channelequalization. Specifically, the routing setup is first set that eachrepeater is configured to know the previous hop source 22. Then,starting from i=1 23, repeaters in the ith hop receive signals y_(i)(m)and calculate w _(i)(m) according to Eq. (5) based on known pilots 24.Then, repeaters in the ith layer set the FIR filters with calculatedwr_(i)(m) 25, before setting i=i+1 26. If i<K 27, then 24-26 arerepeated. Otherwise, the process ends 28. FIG. 14 shows the process ofchannel equalization through feedback. Specifically, the routing setupis first set that each repeater is configured to know the previous hopsource 29. Then, starting from i=1 30, transmitters in the (i−1)th layertransmit orthogonal pilot signals which are known to repeaters in theith layer 31. Then, repeaters in the ith hop receive signals y_(i)(m)and calculate w _(i)(m) according to Eq. (5) based on known pilots 32.Next, repeaters in the ith layer feed back the optimum w ₁(m) to thecorresponding repeater in the (i−1)th layer 33, before setting i=i+1 34.If i<K 35, then 31-34 are repeated. Otherwise, the process ends 35.

One embodiment of this invention is that the repeaters on the same layeruse orthogonal codes (such as m-sequence) or spatial division to avoidinterferences to the repeaters in the next layer. Another embodiment ofthis invention is that these pilots are transmitted in the system GuardPeriod (GP) for a TDD LTE system. In a LTE system, there are somededicated OFDM symbols reserved for pilot transmission which can be usedfor filter coefficients calculations. The upper layer controls thesignal propagation process, and then each repeater knows the previoushop sources, and their pilot signals. With orthogonal pilot sequence,each repeater in the ith layer only receives the signal from the desiredtransmitter in the (i−1)th layer. In addition to the code division,spatial division can also be used to avoid interference. Thetransmitters on some layer that are sufficiently separated in distancecan be scheduled to use the same pilots, e.g., using in the samefrequency and/or code at the same time to avoid interference. Thetransmitters on the same layer can also use high-directional antennas(common for mmWave systems) to send signals to different receivers withsufficient angular separations to avoid interference.

If the direct channel equalization is used, the repeaters in the firstlayer equalize the channels between them and the BS in the downlinkscenario. However, in the uplink scenario, the first layer repeatersequalize the channels between them and the UEs. The channels betweenrepeaters and UEs always have less coherence time than the channelsbetween repeaters. Therefore, it is desired to use the equalizationthrough feedback in the uplink scenario, so that the repeaters in thefirst layer equalize the channels between the repeaters in the firstlayer and repeaters in the second layer, and the repeaters in the lastlayer equalize the channel between them and the BS. This methodguarantees that all the repeaters equalize the channels with longcoherence time, to reduce system resource on equalization. Afterequalization, the total channel between the BS and the UE are still notperfectly flat, because the equalization at repeaters is not perfect andthe channels between the UE and repeaters are not equalized. Oneembodiment of this invention is that the BS or the UE uses OFDM or othermethods to estimate channels. For example, the system has a bandwidth of2 GHz, and the channel is not flat over the 2 GHz bandwidth. However, ifthe 2 GHz bandwidth is divided into W subcarriers based on the OFDMtechnique, then the channel can be considered to be flat for every wsubcarriers. Hence, the UE or the BS can estimate the channel for eachgroup of w subcarriers. In this way, the UE or the BS has a good channelestimation in the overall 2 GHz channel.

The repeaters can achieve equalization using either one of the followingtwo embodiments: (1) Direct amplify-and-forward mode: The FIR filter isconstructed with tap delay lines, and each tap has one or moreadjustable attenuators and/or phase shifters with values set to matchthe values of w _(i)(m); or (2) Sample-and-forward mode: The repeaterdown-converted signals and obtain time-domain samples with anAnalog-to-Digital Converter (ADC), then, the digital signals are passedthrough digital FIR filters, and the output of the filters are thenconverted to analog signals which are up-converted and sent out throughrepeater transmitters.

When all repeaters have obtained the optimum FIR settings, thecommunication between the BS and the UE is the same as withoutrepeaters, since the repeaters operate in the amplify-and-forward mode.However, since the repeaters' inside paths might be asymmetric, specialattention should be paid if the channel reciprocity is used to thedownlink channel estimation. In summary, if the uplink and downlinkchannels of repeaters' inside symmetric, the FIR filters for the uplinkand downlink have the same setting, then the overall channel from the BSto the UE is symmetric. If the uplink and downlink channels ofrepeaters' inside AFR paths are asymmetric, the channel estimation fromthe BS to the UE can be obtained through feedback.

Although the foregoing descriptions of the preferred embodiments of thepresent inventions have shown, described, or illustrated the fundamentalnovel features or principles of the inventions, it is understood thatvarious omissions, substitutions, and changes in the form of the detailof the methods, elements or apparatuses as illustrated, as well as theuses thereof, may be made by those skilled in the art without departingfrom the spirit of the present inventions. Hence, the scope of thepresent inventions should not be limited to the foregoing descriptions.Rather, the principles of the inventions may be applied to a wide rangeof methods, systems, and apparatuses, to achieve the advantagesdescribed herein and to achieve other advantages or to satisfy otherobjectives as well.

REFERENCES

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What we claim are:
 1. A MIMO wireless communication system comprisingone or more BS, one or more repeaters, and one or more UEs, wherein arepeater estimates the channel between itself and its uppercommunication node in the system, computes an equalization filter orequalization coefficients based on the estimation of the channel, andapplies the equalization filter to improve the condition of thecommunication channel containing the one or more repeaters.
 2. Thesystem in claimed 1 wherein improving the condition of the communicationchannel comprises increasing the coherence bandwidth of thecommunication channel.
 3. The system in claimed 1 wherein improving thecondition of the communication channel comprises reducing the delayspread of the communication channel.
 4. The system in claimed 1 furthercomprising that a plural of repeaters are placed in the coverage of a BSso that they forms a network that may contain more than one layer ofcommunication nodes, wherein each repeater receives signals from itsupper nodes that is either a BS or one or more repeaters and transmitssignals to the lower nodes that is either one or more repeaters or oneor more UEs in the downlink transmission.
 5. The system claimed in 1further comprising that the equalization of the channel between twoneighboring nodes is implemented by either the upper node or the lowernode, which is predefined by the system or informed through controlinformation.
 6. The system in claim 1 further comprising that a uppernode transmits a pilot to the lower node and the lower node computes theequalization coefficients with the sampled received signals after theADC, wherein the equalization coefficients are defined as a vectorcontaining multiple complex-valued numbers.
 7. The system in claim 6further comprising that the pilot signals are transmitted in the GP of aTDD wireless communication system.
 8. The system in claim 6 furthercomprising that some OFDM symbols is reserved for pilot transmission forequalization coefficient estimation, wherein the whole OFDM symbol isused for pilot sequence transmission when an OFDM symbol is reserved forpilot transmission.
 9. The system in claim 1 wherein computing anequalization filter comprising computing equalization coefficients assolving an MMSE type problem and the solution has a form of a Wienerfilter.
 10. The system in claim 1 further comprising that the lowernodes feed back the received pilot signal after ADC or the estimatedequalization coefficients to the upper node for equalization coefficientcomputation, wherein the equalization coefficients are defined as avector containing multiple complex-valued numbers.
 11. The system inclaim 1 further comprising that orthogonal pilot sequences is employedby multiple nodes transmitting pilot signals in the downlink forequalization coefficients estimation simultaneously.
 12. The system inclaim 1 further comprising that the same pilot sequence(s) are shared bytwo or more nodes that are sufficiently separated in distance or areusing highly direction antennas with sufficient angular separation. 13.The system in claim 1 further comprising that the transmitter sidebeamforming is applied when the number of antennas on the transmitter,which can be a BS, a UE, or a repeater, is larger than the number ofantennas on the receiving repeater, before the channel equalizationprocess, where the beamforming matrix is calculated based on the channelestimation acquired by the transmitter transmitting pilots to thereceiver and the receiver feeding back the channel estimates, or thereceiver transmitting pilots to the transmitter if channel reciprocityis valid.
 14. The system in claim 1 further comprising that the receiverside beamforming is applied when the number of antennas on the receivingrepeater is equal or larger than the number of antennas on thetransmitter, which can be a BS, a UE, or a repeater, before the channelequalization process, where the beamforming matrix is calculated basedon the channel estimation acquired by the transmitter transmittingpilots to the receiver.
 15. The system in claim 1 further comprisingthat the communication node that computes the equalization coefficientsapplies the coefficients to the received RF signals when it works in theamplify-and-forward or the sample-and-forward mode in the downlink. 16.The system in claim 1 further comprising that the communication nodethat computes the equalization coefficients applies the coefficients tothe transmitting RF signals when it works in the amplify-and-forward orthe sample-and-forward mode in the uplink.
 17. A repeater for using in aMIMO wireless communication system comprising a module for estimatingthe channel between itself and its upper communication node in thesystem, a module for computing an equalization filter based on theestimation of the channel, and a module for applying the equalizationfilter to improve the condition of the communication channel containingthe repeater, wherein the MIMO wireless communication system comprisesone or more BS, one or more repeaters, and one or more UEs.
 18. Therepeater in claimed 17 wherein improving the condition of thecommunication channel comprises increasing the coherence bandwidth ofthe communication channel.
 19. The repeater in claimed 17 whereinimproving the condition of the communication channel comprises reducingthe delay spread of the communication channel.